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DUCBZ/D Thyristors 
Solid State Application Note 

^^^^ AN-4242 

A Review of Ttiyrtstor Characteristic and Applications 

by T.C. McNulty 

niyristors, both SCR's and triacs, are now widely accepted 

in power-control uppliciitions. Willi the cmplutsis in sikH 
applicalions placed on Inw Liist, siimll |i;n.k:iL;i- w/c, iinJ 
circuit simplicity, thyristors satisfy these requirements with 
reliability exceeding that of electromechanical counterparts. 
This Note describes the operation, ratings, characteristics, 
and typical applications of these devices. 

Types of Thyristors 

Thyristors are semiconductor devices that have character- 
istics similar to those of thyratron txtixs; more specifically, 
they are semiconductor switches whose bistable state 
depends on the regenerative feedback associated with a 
p-n-p-n structure. Basically, this group includes any bistable 
semiconductor device that has three or more junclions (i.e., 
four or more semiconductor layers) and can be switched 
from a hi^-impedance (OFF) stale to a conducting (ON) 
sute, and from the condiictii^ (ON) state to the high- 
impedance (OFF) state, within at least one quadrant of the 
principal -volt age characteristics. 

There are several types of thyristors, which differ primarily 
in number of electrode terminals and operating character- 
istics associated with the third quadrant (negative) of the 
vol tagie-cunent characteristics. Reverse^ilocking triode 
tfiyristors, commonly called silicon ccmtroUed rectifiers 
(SCR's). and bidirectional triode thyristors, referred to as 
triacs, are the int)st popular types. Silicon controlled 
rectifiers have satisfied the requirements of many power- 
swilchins; LipplKJlMni^ with inycli greater icliabilily than 
electromechanical or tube counterparts. As the use ol SCR's 

in power applications increased, the need for complete ac 

L-onlnil became apparent. The new family of Ihyristt>r 
^lcvi^,\,'^ fjcncriitcd io pnivitlc bitiircclioiial curreni properties 
is relerrcd lo as triacs. A Iriac can he considered as two 
parallel SCR's (p-n-p-ii) oriented in opposite directions to 
provide symmetrical bidirectional characteristics. 

Two-Transistor Analogy 

The bistable action of thyristors can be explained by analysis 
uf the structure of an SCR. lliis analysis can be related to 
either operating quadrant of a triac because a triac is 
essentially two parallel SCR's oriented in opposite directions. 
A iwo-lransislor analogy of an SCR is illusiraled in Fig. I. 
Fig. shows the schematic symbol lor an SCR, and Fig. 
1(b) shows the p-n-p-n structure the symbol represents. In 
the Iwo-transistor model for the SCR shown in Fig. 1(c), the 
Intercormections of the two (ran^stors are such that regen- 
eratWe action can occur when a proper gate signal is apj^ed 
to the base of the lower n*p-n tranastor. 

In the diagram of Fiy. 2. ilic cniiiier of the upper (p-n-p) 
transistor is returned to [he positive terminal of a dc supply 
through a limiting resistor R^. and ilie emitter of tlie lower 
(n-p-n) transistor is returned lo the negative terminal of the 
dc supply to provide a complete electrical path. When the 
model is in the OFF state, the initial principal -current flow is 
zero. If a positive puise is then applied to the base of the 
n-p-n transistor, the transistor turns on and forces the 
collector (which is also the base of the p-n-p transistor) to a 
low potential; as a result, turreiU (l.|l begins lo tlow. 
Because the p-n-p transistor is then in the active state, 

Trad«TMrk(St R80'*t«"«d'6' IntufmariLin lu'liistied by RCA istjeHmd lo be accuraie and pflnMri in 1IR&I11 77 

Marcs(S) Ragistradats) 

el'atilf. However. t)o responsibililv is assumed by RCA tof 
jU use, nor lor iny mff ingemenrs of patents oi other nghis of 
third parties which nay reoili from its use. No licsue it 
granted by implication or otham^ uikIb any patwii or 
pttaot rigbn of RCA. 


collector current (Ic]-'b2) flow^ ^'■■^'^'^ ""P"" 

transistor and sets up the conditions loi rcgciuTiiiiuii. II the 
cxtcrmi gyle drive is removed, the model remains in (lie ON 
stale as ;i result of the division of currents associated with the 
two transistors, provided that sufficient principal current (1^) 
is available. 



Fig. 1 - Tvm-owi^it(H' analogy of an SCR: (a) schematic 
symbol of SCR; (b) p-n-p-n structure represented by schema- 
tic symbol; fc) two-transistor model of SCR. 

Theoretically, the mode) ^own in Fig. 2 remains in the ON 
state until the principal current flow is reduced to zero. 
Actually, turn-off occurs at some value of current greater 
than zero. This effect can be explained by observation of the 
division of currents as the value of the limiting resistor is 
gradually increased. As the principal current is gradually 
reduced to the zero current level, the division of currents 
within the model can no longer sustain the required 
regencriilion and the nmdel reverts lo the blocking slate. 

The two-transistor model illustrates three features of thyris- 
tors: (1) a gate trigger current is required to initiate 
regeneration, (2) a minimum principal current (referred to as 
"latching current") must be available to sustain regeneration, 
and (3) reduction of principal-current flow results in turn-off 
at some level of current flow (referred to as *1ioldii^ 
otrrent*^ ^SgMly ffesktat than zero. 


Fig.2 - Two-traraiseot fnodBi connected to show a comp/ete 
electrica/ pa^. 


Fig. 3 illustrates the effects on latching and holding current 
for resistive icrniinalion Lit the base of the n-p-n transis- 
tor. The collector current tliroiijili the p-n-p transistor must 
be increased to supply both the base current for the n-p-n 
transistor and the shunt current through the terminating 
resistor. Because the principal-current flow must be increased 
to supply this increased collector current, latching and 
holding current requirements also increase. The use of the 
two-lransistor model provides a more concise meaning lo the 
mechanics of thyristors. In thyrisior fabrication, it is 
generally good practice to use a low-beta p-n-p unit and to 
include internal resistance termination for the base of the 
n-p-n unit. Termination of the n-p-n unit provides immunity 
from '*faise" (non-g^ted) tura-on, and the use of the low-beta 
p-n^ units permits a wider base r^on to be used to support 
the high voltage encount^ed in thyristor applications. 


Fig.3 - Two-transistor model of SCR mth resistive terrrnna- 

tion of the n-p-n transistor base. 

Voltage and Temperature Ratings 

The effects of lcni|feratuic and voltage are important in 
thyristors because these devices possess regenerative action 
and are required to support h^ voltage in the OFF state. In 
the two-transistor model shown in Fig. 2, an increase in 
temperature causes a leakage current which, if allowed to 
migrate to the base of the n-p-n transistors, forcK the 
transistor into the active region. Regenerative action then 
calls for additional leakage cLirreni. and causes the model to 
switch into the ON state and establish a principal -cur rent 
flow. For reliable operation at high temperature, the base of 
the n-p-n transistor ^ouid be terminated with a low value of 
resistance to prevent turn-on as a lesult of hi^temperatun 

Because gate termination is required on all thyristors, RCA 
devices contain a diffused internal gate-cathode resistor (the 
so-caUed '*^rted-emitter" design) and do not require 
external gate termination. Therefore, it is not necessary to 
specify an OFF-state rating under the conditions of external 
gate-resistance termination. The use of this internal shunt 
resistance improves the OFF-state blocking capability, 
provides increased immunity against false ttrm-on, and 
slightly increases gate-current requirements. 


OFF-statc voltage ratings of thyristors are specified for both 
steady-state and transient operation for both forward 

(positive) and reverse (negative) blocking conditions at the 
maximuin junction temperature. For SCR's, voltages are 
considered to be forward (positive) when tlie anode is at a 
positive potential with reference to the cathode. Negative 
voltages are referred to as reverse-blocidng voltages. For 
triacs, voltages are considered to be positive when main 
terminal 2 is at a positive potential with reference to main 
terminal 1; this condition is referred to as first-quadrant (!) 
operation. Third -quadrant (III) operation occurs when main 
terminal 2 is at a negative potential with reference to main 
terminal I . Fig. 4 sliows the principal voltage-current 
characteristics for both SCR's and triacs. 

voltage. When the SCR is in the ON state, the forward 
current is lunited primarily by the impedance of the external 
circuit. Increases in forward (pruicii^) current are acc(nn- 
panied by only a slight change in ON^ate wlta^. 

if the iriac is considered as two parallel SCR's oriented in 
opposite directions to provide symmetrical current flow, the 
tehavior of a triac under poatlve or reverse voltage operation 
is essentially the same as that of an SCR in the forward- 
blocking mode. 

Gate Characteristics 

The breakover voltnge dI a iliyristur can be varied, or 
controlled, by injection of a signal at the gate terminal. Fig. S 


' ffilEAKOVER 





Fig.4 - Principal voltage-current characteristics of SCR's and 

Opa^tion of an SCR under revem-blocjdng voltage is similar 
to that of a reverse-biased silicon rectifier or other semicon- 
ductor diodes. In this operating mode, the SCR exl^bits a 
very high internal impedance, and a small reverse current 
flows through the p-n-p-n structure until the reverse break- 
down voltage is readied, at which time the reverse current 
increases rapidly. For forward (positive) operation, the SCR 
is electrically bistable and exhibits either high impedance 
(forward-blocking or OFF state) or low impedance (forward- 
conducting or ON state). In the forward-blocking state, a 
small leakage current, considered to be of approxhnately the 
same value as that for reverse leakage, flows through the 
p-n-p-n structure. As the forwaid voltage is hicreased, a 
"breakdown" point is reached at which the forward current 
increases i^ndty and the voltage across the SCR decreases 
aln^ptly to a very low voltage, referred to as the forward ON 

Ig4 1,3 1,2 lg|=0 


Fig.5 ' Thyristor breakover as a function of gate current 

^ows curves of breakover as a function of gate current for 
first-quadrant operation of an SCR. A similar set of curves 
can be drawn for both the first and the third quadrant to 
represent triac operation. 

When the gate current Ig is zero, the applied voltage must 
reach the breakover voltage of the SCR or triac before 
switching occurs. As the value of gate current is increased, 
however, the ability of a thyristor to support applied voltage 
is reduced and there is a certain ralue of gate current at 
which the behavior of the thyristor closely resembles that of 
a rectifier. Because thyristor turn-on, as a result of exceeding 
the breakover voltacc. can produce instantaneous power 
dissipLillon during suilcliinj; irjiisilidn. an ineverb.ible 
condition may exist unless the magnitude and rate of rise of 
principal current is restricted to tolerable levels. For normal 
operation, therefore, thyristors are operated at applied 
voltages lower than the breakover voltage, and are made to 
switch to the ON state by gate signals of sufficient ampUtude 
to assure complete turn-on independent of the appUed 
voltage. Once the thyristor Is triggered to the ON state, the 
principiii-current flow is independent of gate voltage or gate 
current, litkI the device remains in the ON state until the 
principal-cuireiii How is reduced lo a value below the holding 
current required to sustain regeneration. 

The gale voltage and current required to switch a thyristor 
from its high-impedance (OFF) stale tn its low-impedance 
(ON) state at maximum rated forward anode current can be 


^termined from the circuit ^own in Bg. 6. Re^or R2 is 
selected so that the anode current specifled in the manufac- 
turer's ratings flows when the device latches into fts 
low-impedance or ON state. The value of Ri is gradually 

decreased iiiitil ilie device under test is switched from its 
OFF stale lo iis !o\v-imped;ince or ON state. The values o! 
gate current and gate voltage immediately prior to switching 
are the values required to tr^er the thyrtstor. For an SCR, 
tl»re is only one mode of gate firing capable of switching the 
device into the ON slate, i.e., a positive gate signal for a 
positive anode voltage. If the gate polarity is reversed 
(negative voltage), the reverse current flow is limited by the 
value ni l< J ami ilie gaie-catliude internal shunt. The value of 
power dissipated lor the reverse gate polarity is restricted to 
the maximum power-dis^paticm limit impcoed by the man- 

j-a;x-0— |- 

12V V 

T L 


Fig.6 - Circuit used to measure thyrtstor gate voltage and 
current switching threshold. 

Because of its complex structure, a triac can be triggered by 
either a positive or a negative gate si^al regardless of the 
voltage polarity across the main terminals of the device. Fig. 
7 illustrates the triggering mechanism and current flow 
within a triac. The gate trigger polarity is always referenced 




IT 2 


(c) m (*) 

Fig.7 • Cio'fent flow in a triac. 

to main terminal 1 . Tlie potential difference between the two 
terminals is such that gate current flows in the direction 
indicated by the dotted arrow. The polarity symbol at main 
terminal 2 is also referenced to main terminal 1. The 
semiconductor materials between the various junctions wi&- 
in ilie pellet are labeled "p" and "n" to indicate the type of 
majority-carrier concentrations within the material. 

For the various operating modes, the polarity of the voltage 
on main terminal 2 with respect to main terminal I is given 
by the quadrant in which the triac operates (either I or III), 
and the polarity of the gate signal used to trigger the device is 
given by the proper symbol next to the operating quadrant. 
For the I(+) operating mode, main terminal 2 and the gate 
are both positive with respect to main terminal 1 . Initial gate 
current flows into the gate terminal, through the p-type 
layer, across the junction into the n-type layer, and out main 
terminal 1 , as shown by the dotted arrow. As gate current 
flows, current inultipiication occurs and the regenerative 
action u/ilhin the pellet switches ihe triac to its ON slate. 
Because of the polarities indicated between the main 
terminals, the principal current flows through the p-n-p-n 
structure as shown by the soUd arrow. Similarly, for the 
other three op^atii^ modes, the initial gate-current flow is 
^own by the dotted arrow, and principal-cunent flow 
through the main terminal is shown by the solid arrow. 

Because the direction of principal current influences the ^te 
trigger current, the magnitude of the current required to 
trigger the triac differs for each mode. The operating modes 
in which the principal current is in the same direction as the 
gale Liirrem require less gate trigger current: modes in which 
the principal current is in opposition to the gate current 
require more gate trigger current. 

Because triacs are bidirectional, they can provide full-cycle 
(360-degree) control of ac power from either a positive or a 
negative gate-drive signal. This feature is an advantage when 
it is necessary to control ac power from low-level logic 
^sterns such as inte^ated-circuit logic. With gate-power 
requirements for turn-on in the milliwatt r^on, triaps are 
capable of controlling power levels up to 10 kilowatts. Thus, 
the power ^in asscx:iftted m&i these thyristors far exceeds 
that of tranalstcff counterparts in the saniconductor switch- 
ing field. 

Like many other semiconductor-device parameters, the mag- 
nitude of gale trigger current and voltage varies with the 
junction temperature. As thermal excitation of carriers 
within the semiconductor material increases, the increase in 
leakage current makes it easier for the device to be tri^red 
by a gate signal. Therefore, the gate becomes more sensitive 
in all operating modes as the junction temperature increases. 
Conversely, if a triac or SCR is to be operated at low 
temperatures, sufficient gate trigger current nuisi he provided 
to assure triggering of all devices at Ihe luwest operating 
temperature expected in any particular application. Varia- 
tions of gate-tri^er requirements are ^ven in the p^U^d 
data for individual thyristors. 


The ^te current specified in published data for thyristors is 
the dc gate trigger current required to switch an SCR or triac 
into its low-impedance state. For practical purposes, this dc 
value can be considered equivaleni (o a pulse current that has 
a minimuni pulse width nf 50 inicrosccnnds. For gate-current 
pulse widths smaller than 50 microseconds, the pulse-current 
curves as»:jciated with a ^iticu;kr d«me ^ould be used to 
assure tura^. 

When pulse triggering of a thyristor is required, it is always 
advantageous to provide a gate-current pulse that has a 
magnitude exceeding tlie dc value required to trigger the 
device. Tlie use ot large trigger currents reduces variations in 
turn-on time, increases di/dt capability, minimizes the effect 
of temperature variation on tri^ering characteristics, and 
makes possible very short switching times. When a thyristor 
is initially triggered into conduction, the current is confined 
to a small area which is usually tlic more sensitive part of the 
cathode, li' I he anode cm rem magnitude is great, the 
localized instantaneous power dissipation may result in 
irreversible damage unless the rate of rise of principal current 
is restricted to tolerable levels to allow time for current 
^reading over a larger area. When a much larger gate signal is 
applied, a greater part of the cathode is turned on initially; as 
a result, turn-on time is reduced, and tlie thyristor can 
support a much larger peak anode inru^ current. 

Switching CtiaraKAeris^ 

Ratings of thyristors. are biased upon the amount of lieat 
generated within the device pellet and the ability of the 
device package to transfer the internal heat to the external 
case. For high-performance applicaliiMis in which switching 
of high peak current values but narrow pulse widths is 
desired, the internal ener^ dissipated during the turn-on 
process must be determined to assure that power dissipation 
is within ratings. 

When thyristors (either triacs or SCR's) are triggered by a 
gate signal, the turn-on time consists of two stages, a delay 
time td and a rise time tf. as shown in Fig. 8. The total 
turn-on time tgt is defined as the time interval between the 
initiation of the gate signal and the time for the principal 
anode current flow through the thyristor to reach 90 per 
cent of its maximum value for a resistive load. The delay 
lime I J is defined as the time interval between the 
5()-pci-coni |iniiii •>] \Uv' lc;uliiig edge oC the gate trigger 
voltage and the lU-per-cent point uf the principal current for 
a resistive load. The rise time is the time interval required 
for the principal current to rise from 10 to 90 per cent of its 
maximum value. The total tumoB time ton the sum of 
both delay and rise time (tj + tf). 

Althougli IliL- ihyrisior is alTecied to SOrae extent by the 
.peak off-slate voltage and the peak on-state current level, the 
tuiH'On time is influenced primarily by the magnitude of the 
j^tfiTtrigger pulse current, as shown in Fig. 9. Faster turn-on 
tiine for ^gSF s^U drive is a emilt of a decrease in ^teiy 

F/g.8 - Waveshapes illustrating thyristor turn-on time. 

time associated with the thyristor because of the increased 
current density at the gate-cathode periphery. Of major 
importance in the turn-on time interval is the relationship 
between thyristor voltage and principal current flow through 
the thyristor. During the turn-on interval, the dynamic 
voltage drop is high and the current density can produce 
localized iiot spots in the pellet area. Thereiore. ii is 
important that power dissipation during turn-on be restricted 
to levels within device specifications. 


FigS - Thyristor turn-on time as a funption of sote 0-igger 

Turn-off time of a thyristor can be associated only with 
SCR's. In triacs. a reverse voltage caniini hi.' used to prLivide 
circuit-commutated turn-off voltage because a reverse voltage 
ai^Jicd to ooe h^f of the jt^mc Mm^tm^ vmM be a 


forward-bias voltage to the other half. For tuxn-off times in 
m SCR, the recovery period cmskts of two stages, a revere 
recovery time mi a gate reiaovery time, as shown in Fi^. 10. 

toff ^ 


Fig. 10 - Waveshapes illustrating thyristor turn-off time. 

When ihe loi ward current of an SCR is reduced to zero at 
the end of a conduction period, application of reverse vohage 
between the anode and cathode terminals causes reverse 
current to flow in the SCR until the time that the reverse 
current passes Its peak value to a steady-state level called the 
reverse recovery time tfr. A second recovery period, called 
the gate recovery time, tgf, must fheii elapse lor the 
forward-blocking junction to establish a depletion region so 
that forward-blocking voltage can be reapplied and success- 
fully blocked by the SCR. The gate recovery time of an SCR 
is usually much longer than the reverse recovery time. The 
total time from the instant reverse recovery current begins to 
flow to the start of the forward- blocking voJtage is referred 
to as circuit-commutated turn-off time tq. 

Turn-off time depends upon a number of circuit parameters, 
including on-state current prior to turn-off, rate of change of 
current during the forward-to-rever^ tran^tion, reverse- 
blocking voltage, rate of change of reapplied forward voltage, 

gate trigger level, the gate bias, and junction temperature. 
Junction temperature and on-state current have a more 
significant effect on turn-off than any of the other factors. 
With turn-off time specified on the manufacturer's data sheet 
and dependent upon the conditions as outlined above, 
turn-off time specifiMtion is only meaningful if all of the 
above critical parameters are airailable in the actual applica- 

For applications in which an SCR is used to control 60-Hz ac 
power, the entire negative half of the sine wave is a turn-off 
condition and more than adequate for complete turn-off. For 
ai^catas in wbK^ ft« is used to eoMVol t^e (pKj^t 


of a full-wave rectifier bridge, however, there is no reverse 
voltage available for turn-off, and complete turn-off can be 
accomplished only if the bridge output is reduced to zero 
volts or the principal current is reduced to a value lower than 
the device holding current. 

Because turn-off times are not associated with tr^s due to 
the physical structure of the device, a new term is introduced 
called "critical rate of rise of commutation voltage"\ or the 
ability of a triac to conmiutate a fixed value of cun ent under 
specified conditions. The rating can be explained by con- 
^deration of two SCR's in an mverse parallel mode, as shown 
in Fig. 1 1. SCR-1 is assumed to be in the conducting stale 

Fig. 1 1 - Circuit wsetf to d&nonsiram critical raw of rise of 
commutation vot^^s. 

with forward current established. As the principal current 
flow cros^ the zero reference point, a small reverse current 
flows in SCR-1 until the time that the SCR reverts to the 
OFF state. The principal current is then diverted to SCR-2, 
provided that sufficiMit gate current is available to that 

The structure of a triac shown in Fig. 12 indicates that the 
main blocking junctions are common to both halves of the 


I 1 

I P 

. I 








Fig. 12 - Structure of a triac. 

device. When ihc Hrst half of the triac structure (SCR-1) is in 
Ihe conducting state, a quantity of charge accumulates in the 
n-type region as a result of the principal current flow. As the 
^Inc^pfli euff^t Crosse the zero t^^^r^^ pn^U a smaH 


reverse current is eslahlished as a resuil (if (ho -.■liaree 
remaining in [he n-lypc region. BcL;insc llio n-upL' region i.s 
onimon to both halves of the devices, this reverse recovery 
current becomes a forward current to the second half of the 
triac. The current resulting from stored charge may cause the 
second half of the triac to go into the conducting state in the 
absence of a gate signal. Once curr^t conduction has been 
established by application of a gate signal, therefore, com- 
I^ete loss in power control can occur as a result of 
interaction within the n-type base region of the triac unless 
sufficiuMU time elapses to assure turn-off. It is imperative that 
triac manufacturers provide sufficient information regarding 
^ — commutating capability und» maximum current and 
case-temperature conditions so that triac control of ac power 
for resistive loading in a 60-Hz power source can be assured. 

Commutation of triacs is more severe with inductive loads 
than with lo.^istivi.' londs Iuvmu^o of the plnise !;ig between 
voltage and current associated with inductive loads. Fig. 13 
sh&ms the VKawforms for an inductive load witii If^fig 

Fig. 13 - Waveshapes of commu^ting dv/dt characteristics. 

current power factor. At the time the current reaches zero 
crossover (point A), the half of the triac in conduction begins 
to commutate when the principal current falls below the 
holding current required to sustain regeneration. Because the 
high-voltage junction is common to both halves of the triac, 
the stored charge can be neutralized only by recombination. 
At the instant the conducting half of the triac turns off. an 
ipplied voltage opposite to the current polarity is applied 
across the Iriac terminals (point B). Because this voltage is a 
forward bias to the second half of the triac, the sudden 
reapplied vohage in conjunction with the remifining stor^ 
charge in the high-voltage junction reduces the over-all device 


L^ipiihility to support a fast rate of rise of applied voltage. 
Mil.' rcsuti is a loss of power control to the load, aTid the 
device remains in the conducting state in absence of a gate 
signal. Therefore, it is imperative that some means be 
provided to restrict the rate of rise of reapplied voltage to a 
value which will permit triac turn»o£f under the conditions of 
inductive load. 

An accepted method for keeping the coinmutaling d\ di 
within tolerable levels during triac lurn-olf Is to use an RC 
snubber network in parallel with the main terminals of the 
triac. Because the rate of rise of applied voltage at the triac 
terminal is a function of iht load impedance and the RC 
snubber network, the circuit can be evaluated under worst- 
case ctmdttions of operating case temperature, ntaximum 
principal current. luiJ ain value of coniunciinn LtiiL^le. The 
values of resistance and capacitance in the siuibbei are then 
adjusted so that the rale of rise of connnutatini; tlv/dl stress 
is witliin the specified minimum limit under any of the 
conditions mentioned above. The value of snubber resistance 
should be high enough to limit the snubber capacitance 
discharge currents during turn-on and dampen the LC 
oscillation during commutation (turn-ofO- Any combination 
of snubber resistance and capacitance that provides the 
requiiemartso^iffiiied sSa&9t is considered satisfactory. 

Some of the factors affecting commutating dv/dt capability 
<^ triacs are temperature, current magnitude, rate of change 
of current during commutation, and frequency of the implied 
principal current. With frequency directly related to commu- 
tating di'dl. early triac use was restrict^ to 60-Hz applica- 
tions. Continued technological advances in triac device struc- 
ture has resulted in faster "turn-off' capability and made 
possible a new family of triacs having 4{K)-Hz commutating 
capability that is now being offered to circuit designers w4io 
must work with 4<K)-Hz source voltages. 

Another important parameter for thyristors is the "critical 
rate of rise of off-state voltage". A source voltage can be 
suddenly applied to an SCR or a triac which is in the OFF 
state through either closure of an ac line switch oi transient 
voltages as a result of an ac line disturbance. If the fast rate 
of rise of the transmit voltage exceeds the device rating, the 
thyristor may switch from the OFF state to the conducting 
state in the absent of a gate rignai. If the thyristor is 
controlling alternatii voltage, "false" turn-on (non-gated) 
resulting from a Iran -nt imposed voltage is limited to no 
more than half the applied voltage because turn-olT occurs 
during the zero current crossing. However, if the source 
voltage suddenly applied to the OFF thyristor is a dc voltage, 
the device may swtch to the ON state and turn-off could 
then be achieved only by circuit interru^^s. The switc^g 
from the OFF state is caused by the inteanal capacitance of 
the thyristor. A steep-rising voltage dv/dt impressed across 
the terminals of a thyristor causes a capacitance-charging 
current to flow through the device. This charing current 
(i=Cdv/dt) is a fu[H;tion of the rate of rise of applied off-state 
volt^. If the rate of rise of voltage exceeds a critical value. 



the capacitance-charging current exceeds the gate trigger 
current and causes device turn-tin. Operatitin at elevated 
junction temperatures reduces the thyristor ability to sup- 
port a steep rising voltage dv/dt because less gate current is 
required for turn-on. The effect of temperature on the 
critical rate of rise of off -state voltage is shown in Fig. 14. 

t S l.Oi 

Vd = VoROi* 


Fig. 14 - Critical rate of fitB Gf aff^m^mfmgt^mafytiethn 

of case temperature. 

Voltage transients which occur in electrical systems as a resuli 
of disturbance on the ac line caused by various sources sucii 
as energizing transformers, load switching, solenoid closure, 
contactors, and the like may generate voltages which are 
above the ratings of thyristoi^ and result in spike voltages 
exceeding the critical rate of rise of off-state voltage 
capability. Thyristors, in general, switch from the OFF state 
to the ON state whenever the breakover voltage of the 
device is exceeded, and energy is then transferred to the 
load. Good practice in the use of thyristors exposed to u 
heavy transient environment is lo provide some form of 
transient suppression. 

For applications in which low-energy, long-duration tran- 
sients may be encountered, it is advisable lo use thyristors 
that have voltage ratings greater than the highest voltage 
transieat expected in the system to provide protection 
against destructive transients. The use of voltage clipping 
cells is also effective. In either case, analysis of the circuit 
application will reveal the extent to which suppression 
should be employed. In an SCR application in which there is 
a possibility of exceeding the reverse-blocking voltage rating, 
it is advisable to add a clip cell or to use an SCR with a 
higher reverse4>lockii^ voltage rating to minimize power 
dissipation in the reverse mode. Because triacs generally 
switch to a low conducting state, if the di/dt buildup of the 
principjd current flow after turn-on is within device ratings it 
is safe to assume that reliable operation will be adiieved 
under the specified conditions. 

The use of an RC snubber is mo^ eff^ve in reducing the 
effects of the high-ene^ short<duration transients more 

frequently encountered in thyristor applications. When an 
RC snubber is added at the thyristor terminals, ihc rate of 
rise of voltage at the terminals is a function of the load 
impedance and the RC values used m the network. In some 
applications, "false" (non-gated) turn-on for even a portion 
of the applied voltage cannot be tolerated, and circuit 
response to voltage transients must be determined. An 
effective means of generating fast-rising transients and 
observing the circuit response to such iransients is shown in 
Fig. 15. This circuit makes use of the "splash" effects of a 
mercury-wetted relay to transfer a capacitor charge to the 
input terminals of a control circuit. This approach permits 
generation of a transient of known magnitude whose rate of 
rise of voltage can easily be displayed on an oscilloscope. For 
a given load condition, the values in the RC snubber network 
can be adjusted so lliat the transient voltage at the device 
terminals is suppressed to a tolerable level. This approach 
affords the circuit designer with nieaninglul inI"orinaiion as 
to how a control circuit will respond in a heavy transient 
erkviTQimieBi circu^ is capable of genexatii^ tcansieBt 

, 5.6lt 

► 1 V\A>- 


Fig. 15 - Circuit used to generate-fast rising transients. 

voltages in excess of 10 kiiovolts per microsecond, which 
exceeds industiijl generated transients. The response of a 
lOO-millihenry solenoid control circuit exposed to a fast- 
rising transient is ^own tn Fig. 16. 

Use of Diacs For Control Triggering 

Basically, thryristors are current-dependent devices, and the 
magnitude of gate current Igt voltage Vgt required to 
trigger a thyristor into the on-state varies. The point at w^ch 
thyristor tri^ring occurs depends not only on the required 
gate current and voltage, but also on the trigger source 
impedance and voltage. Fig. 17 shows a laniily of curves 
representing the gate-circuit load line between (he open- 
circuit source voltage and the short-circuit current for 
different time intervals. In a circuit v^fa applies time- 
dependent variable voltage Vac to a load and the gate trig^r 
current required to tei^a the thyristor is derived from Ae 
same source Vac, devices that have a gate current Igi are 


Fig. 16 ■ Waveforms showing response of a 100-miftffienry 
solenoid control circuit to a fast-rising transient. 

triggered earlier in the ac cycle tiian devices lluil have a 
higher gate trigger current Fig. 3. Although tiic circuit is 
capable of providing variable power to the load, it is heavily 
dependent on the gate current distribution, and results in 
uncontrolled conduction angles for a given value of gate 
series resistance. Furthermore, the circuit does not provide 
the recommended gate-current overdrive for switching of the 
fasl-ris'ag high-amplitude load currents present in resistive 
loading. A more efficient circuit for control of variable 
power to a load that eliminates the need for tight gate- 
current distribution uses a solid-state trigger device, called a 
diac, which is voltage dependent. 

The diac, often referred to as a bidirectional trigger diode, is 
a two-terminal, three^layer, transistctf-rlikc: stmctuie tl^t 

Fig.17 ' Thyristor gate^'reuit load fine for different time 

exhibits a hi^-impedance blocking state up to a breakover 
TOlt^e V(BO)' above which the device enters a negative- 
resistance t^<m. The chmmt&as^c curve in F^. 18 ^ows 



1 J 

^_ ^ 

Fig. 18 - Diac voltage-current characteristic. 

the negative characteristics associated with diacs when liiey 
are exposed to voltages in excess of the breakover voltage 
V(BO)- Because of their bidirectional properties and break- 
over voltage level, diacs axt useful in triac control circuits in 
which variable power is to be supplied to a load. Because of 
their nei;alive characlerislic slope, diacs can also he used with 
caiXKilnis In iiroviJe (he kisi -risiiiy h igli-iiKiyni tudt' trigger 
current pulse i econnnciuled in ihyristor applications which 
require ctTicieni gale itirn-on for the purpose of switching 
iiigli-Ievel load currents. 

in normal applications, diacs are used in conjunction with 
RC phase networks to trigger triacs, as shown in Fig. 19. The 


f^i t0 ^ tfee o/ diac wa'tfl R€ pfiase network to trigger triac. 



RC phase network provides an initial phase-angle displace- 
ment0 so that conduction angles in excess of 90 degrees can 
be realized. As the voltage on the capacitor begins to build 
up in a sinusnida! manner, ihe breakover voltage VfgQ) of 
the diac is rcak.Iied, the trvdc is lurned on, and a portion of 
the ac input voltage is provided to the load, as represented by 
the angle a. As previously mentioned, the diac offers a 
negative-resistance region and is capable of providing current 
pulses whose mi^nitude and pulse width are a function of 
the capacitor C and the combined impedance of the diac and 
the gate and main terminal of the triac. When the vohage on 
the capacitor C reaches the breakover voUage V(gO), the 
capacitor does not discharge completely, but is restricted to 
some fiiiiie level as :t icsiili of" ilic diac negative-impedance 
characteristic at iiigli values of pulse current. Fig. 20 shows 
the peak pulse current of a di^ as a ft^ne^n of 
capacitances of the phasing capacitor C. 

0.04 0.06 

Fig*20 - Peak pulse current of a diac as a function of phasing 

Power Controt Using Th^iston 

In the control of %c fmm by n^^^ of semiconductor 
devices, emphasis has been placed an circuit simplicity, low 
cost, and small over-all package size. Thyrislors meet these 
goals, and are also capable of providing eilher fixed or 
adjustable power to the load. Fixed power is achieved by use 
of the thyristor as an ON-OFF switch, and adjustable power 
through the use of an RC phase network which provides 
variable phase-gating operation. The following section dis- 
cusses both SCR and triac circuit operations, ajid analyas of 
SCR and triac behavior for various circuit conditions. 

Man)' fractional-horsepower motors are series-wound 
'"universal" motors cajiable of operation from cither an ac or 
a dc source. In (he early stages of thyristor control, SCR's 
found wide acceptance in the control of universal motors, 
particularly in the portable power tools market. SCR's are 
capable of providing speed control over half of an ac sine 
wave, and, if full power is rcc|uired, a simple shorting switch 
across the SCR provides the necessary function; such a 
switeh is ^lawn tn Hg. 21, Tias-off pfo^a^ters Jot liiis 

Fig.21 - Simple SCR half-wave control circuit. 

circuit are not critical because the SCR has a half-cycle of 
applied negative voltage in winch in recover. The SCR 
provides a reliable, highly efficieni, long-life control for 
half-wave control circuits. 

Fig. 22 shows a full-wave bridge that feeds a resistive load 
and uses an SCR as the control element for load current. 
Power control is accomplished by SCR turn-on at various 
conduction angles with respect to the applied voltage. The 
criteria for (urn-off in this circuit is impt)rlant because the 
SCR must recover its forward-blocking state during the time 
that the forward current st<^ flowing. Although this time 
interval may appear to be very smafl, close analysis of the 
voltage wave during the transition time in which tiie 
full-wave bridge reverses direction reveals that substantial 
time exists for turn-off. 

Fig.22 - Full -wave SCR bridge circuit. 

Fig. 23 shows one-half of the bridge during the time that the 
forvrard current is approaching zero current. Two diodes are 
in series with the SCR; it is generally accepted that a diode 

Fig.23 - Half of bridge ciwuit of Fig. 22 mhm forward 
eumnf^ireacfm iero for a resiso've load. 


. AN-4242 

voltage of approximately 0.6 volt is required to maintain 
each diode in conduction. If it is further assumed that a 
^voltage of approximately 0.6 volt is required across the SCR 
to maintain conduction, the sum of the voltage drops over 
the circuit requires 1 .8 volts; below this value, the SCR drops 
out of conduction. As the bridge reverses current direction, 
the same analysis holds true, i.e.. forward conduction currcn: 
is not resumed until the sum of the voltage drops exceeds 1 .K 

The waveform during the interval that the voltage wave goes 
from 1 .8 volts to zero can be analyzed by reference to Fig. 
24. A half-cycle (180 degrees) of conduction requires 8.3 
milliseconds, one degree being equal to approximately 46 
microseconds. Because a sine wave is linear for very small 
angles, a graph can be constructed to show the time interval 
during which the voltage is less than 1 .8 volts for various 
magnitudes of applied voltage. Analysis of the voltage wave 
for an angle of one degree shows that an input voltage of 120 
volts rms results in a voltage equal to 2.9 volts, which decays 
to zero in 46 microseconds. Because the SCR is 
non-conducting below a circuit threshold of 1.8 volts, a time 
of 28.5 microseconds then elapses while the voltage decays 
from 1 .8 volts to zero. An equal time is required for the 
bridge to build up to the threshold voltage of 1.8 volts. 
Therefore, a total exposure time of 57 microseconds elapses 
in which the SCR is allowed to r^;ain its forward-biockii^ 

As shown in Fig. 24, increasing the magnitude of the applied 
voltage source to 240 volts rms cuts in half" the time interval 
which the SCR is allowed for turn-otT. Further increases in 
input voltage magnitude result in shorter turn-off periods. 

Fig.24 - Wavefonn of a'rcuit in Fig, 72 as v^tage wave goK 
from 1.8 vo/ts to zero. 

This analysis gives a cleai . wc!I-Lk-rincd pi. lUK' ol ihe iiiin-ofr 
time available lor a rcsisiive load, ilowcvei, I'oi reactive 
loads, such as fractional-horsepower inoK^is, the turn-off 
conditions, including turn-olT lime and dv/dl stress, are more 
difficult to define b«:ause they are affected by a number of 
variable, including the back EMF of the motor, the ratio of 
mdtictance to rMista^ce. the mpt® IpadiQg, and the ghase 
ai^e (tf motor current to source volt^. Normdfy, turn-off 

times for SCR's are industry-standardized to include peak 
forward current, rate of rise of reverse current, peak forward 
blocking voltage applied, and rate of rise of applied blocking 
voltage. The prince of the applied reverse current helps to 
shorten turn-off times because the reverse current sweeps out 
the charge in the blocking junction. Fot SCR operation from 
J tiill-\\j\c birUgc m which iliere ui' .ippie^iable reverse 
vi_li:rj- A.Hhibie. turn-oil is accomplished through recombin- 
aii'MK .Ml. I liio eiTccis of circuit loading m SCR operation 
musl be clearly evaluated. 

Full-wave ac switching can also be performed by use of two 
SCR's in an inverse parallel mode, often referred to as a 
"back -to-back" SCR pair, as shown in Fig. 25. This circuit 
can be used as a simple static switch or as a variable phase 
conir()l circuit. It docs not make use of a full-wave diode 
bridge, but simply uses the SCR's in an alternating mode. 
The circuit has the disadvantage of separate trigger logic, but 
possesses an inherent advantage in higher-frequency applica- 
tions because advantage can be taken of the periods of the 
alternating voltage in which either device may recover to its 
blocking state. During the half-cycle of the applied voltage 
that SCR-1 is conducting, SCR-2 is reverse-biased arui can 
recover its blocking state. Because of the applied reverse 
voltage and associated time of the half<yde volti^, turn-off 
times are not critical. 

Fig.25 - Full-wave ac switching circuits using a "bacf(-to- 
ttac/t" SCR pair. 

This two-SCR circuit is often favored over a triac circuit, 
even tfiotigh separate trigger sources aic rctjinreJ, because it 
is supptiscd to have belter commuiaimg i-apaliiliiv. Fig. 26 
shows the waveforms of commuianng d\ Ji |oi the SCR 
circuit. If the load is inductive with lagging current power 
factor, the conducting SCR commutates at the time the 
principal current reaches zero crossover (point A) and reverts 
to the blocking state; a reapplied voltage of opposite polarity 
equal to the source voltage then appears across the non- 
conducting SCR. Because tliis voltage is a forward-bias 
voltage to the non-conducting SCR, device turn-on can occur 
if the rate of rise of applied forward vohage exceeds the 
device rating for critical rate of rise of off-state voltage. For 
inductive loading in an inverse-parallel-mode SCR applica- 
tion, power control to the load tm lost if the rate of rise 
of applied voltage is exceeded. 



Fig.26 - Waveforms of commutating dv/dt for SCR circuit of 
Fig. 25, 

Although it may appear that the rate of rise is extremely fast, 
doser circuit evaluation reveals that the dv/dt stress is 
restricted to some finite value which is a function of the load 
reactance L and the device capacitance C. Theref*.)re. it is 
important thai the rale of rise of applied voltage during 
commuiaiion noi o\vocd !lio device specillcaiion for critical 
rate of rise of ofl-state voltage under worst-case condition or 
luiFeliable op^ation may result. It is genially good practice 
in inverse-parallel operation to use an RC snubber network 
across the SCR pair to limit the rate of rise to some finite 
value beiow the minimum requirements, not only to limit the 
voltage rise during commutaUon, but also to suppress 
transient voltage that may occur as a result of ac line 

As previously meniioned, ilie use of semiconductor devices 
for ac power control has emphasized circuit simplicity, low 
cost, and small over-all pack^e size. The development of the 
bidirectional triode thyristor, referred to as a triac, adiieved 
all of these goals. Triacs can perform tlw same functions as 
two SCR's for full-wave operation, and also ^idify gate 
logic requirements for triggering. 

A simple, inexpensive triac circuit that can provide variable 
power (o a load over a full cycle of applied voltage is the 
lighl-diininei circuit. This circuit cotitiiins a diac, a liiac. and 
an RC phase-control network. The basic liglil-dininier circuit 
is described below because it provides a good example of 
triac behavior as related to load requirements and of the 
(^«:ation of a dUc in an RC {diase-ctmtool drctiit. 

Fig. 27 shows the basic triac-diac li^t-dimmer control drcuit 
with the triac connected in series with the load. During the 
beginning of each half-cycle, the triac is in the off-state and 
the entire line voltage is across the triac; therefore, n 
voltage appears across the load. (Actually, there is some 
voltage across the load as a result of triac leakage currents, 
which are a function of applied voltage and junction tempera- 
ture. However, these l^tage cunents are lelativety small, at 
most in the ndlfiamp^ xangie, and the resulting load volta^ 
are generally ignored.) 

The RC charge-control circiiii is in parallel with the control 
triac. and the applied voltage serves to charge the timing 
capacitor C through the variable resistor R. When the voltage 
across C reaches the breakover voltage V(bO) of the diac, tht 
capacitor discharges throi^ the diac and the gate-to-main- 
terminal-1 impedance of the triac and turns on the control 
triac. At this point, the line voltage is transferred to the load 
for the rcniainvior of ilic Ltpplicd half-cycle vollage. As the 
load current reverses direction (zero crossing), llie Iriac turns 
off and reverts to the blocking state. This sequence of events 
is repeated for every following half-cycle of applied voltage. 

Fig.27 ■ Basic triac-diac light-dimmer control circuit. 

If the value of resistance R is decreas«i, the capacitor charges 
to the breakover voltage V(bo) of t^tac earlier in the ac 
cycle; the power supplied to the load is then increased and 
the lamp intensity is effectively increased. If the value of 
resistance R is increased, triac tr^ering occurs later in the ac 
cycle and applied volt^e to the load is reduced; the result i 
decreased lamp intensity. Therefore, changes in the resistant 
value R effectively apply variable power to a load (which is a 
lamp load in the circuit of Fig. 27, but could also be a motor 
load or heating element). 

Although the load is arbitrarily placed m series with main 
terminal 2, the circuit performs equally as well if the load is 
shifted to main terminal 1. (Actually, any commercial lamp 
dimmer available has two wires brought out for external 

connection, and the chance thai the load will be connected 
to main terminal 1 is 50 per cent.) 'flic only icquircments fo'" 
reliable operation are that Ihc RC phase network be i 
parallel with the triac and that capaciioj discharize loop 
currents be directed from the diac to the triac gate and main 
terminai 1. Alta^ the baac l^^t-contidi circuit operates 


with the component arrangement shown in Fig. 11. addi- 
tional Loniponcnts arc ot'lcii aiklcd (n rediKi' hvslercsis 
I'fecls, extend the eft'ective range of power control, and 
<$upprmnt^4m|aeficy interference. 

Hysteresis in triac phase-control circuits is referred to as the 
ratio of applied loyd vohage when Ihe triac initially turns on 
(as control poientiomeler is slowly reduced frt)ni some high 
vylue) lo tiie value of load vohage prior to "cxtinguishinj;" 
(as the control potentiometer is slowly increased lo some 
h^er value). If the circuit has high hysteresis, the control 
potentiometer travel may be as high as 25 per cent before 
triac turn-on occurs, after which the control potentiometer 
may be turned back IS pwr cent before the triac "ex- 
tinguishes". Hysteresis is an undesirable feature if the circuit 
application requires low-level lamp iliuminaiion hcc;Hisc a 
momemtary drop in line vnliage may resuli in the triac 
"extinguishing" or missing one half-cycle ol' applied vt)liage 
when the capacitor voltage is barely equal lo ihc breakover 
voltage V^BO) ^ condition exists, the 

control potentiometer must be reduced to "start up" the 
triac again. 

Hysteresis is a result of the capacitor discharging through ihe 
diac and not recovering the original voliage prior lo 
trig^ring. Fig. 28 shows the waveforms of the charging 

Fig.^ - Chargb^ cycle of capacitoMiiac netvvoek in Fig. 27 
(high hystavsh). 

capacitor C as rebted to the applied line voltage. The initial 
displacement angle is a result of the phase angje due to the 
value of the RC components used. As the value of the 
control potentiometer is slowly reduced, the value of 
charging voltage reaches the breakover voltage V([jq) of Ihe 
diac. and the iriac allows thai portion ol' the ac wave 
remaining to appear at the load, as represented by the shaded 
area at the first trigger point. At this point, there is an abrupt 
change in capacitor voltage (A V). Therefore, as the capacitor 
charge reverses direction, the second trigger point is reached 
much earlier in the next half-cycle, and that portion of the ac 
wave remaining appears across Ihe load, as represented by ihe 
sliadeii area at the second trigger poinl. The second trigger 
point and subsequent trigger points represent the steady-state 
level at which triggering occurs. Some reduction in hystereas 
can be realized by inserting a redstor in series with the diac 


to reduce the effective diac negative resistance and minimize 
the change in capacitor vohage. However, this change reduces 
the gate current pulse and, if not carefully controlled, may 
result in di/dt failures because the triac switches l^h- 
magnitude current under minimum gate drive. 

A more effective method of reducing hysteresis is to use a 
second RC time constant, or a "dotdtle-tkne-constant** 
circuit such as that diown in Bg. 29. As C2 si^^ tfae 

Fig.29 • "Doub/e-time<onstant" i/ght-control circuit. 

charging voltage for the diac breakover voltage V(BO)' the 
abrupt change is capacitor voltage during diac turn-cm is 
partially restored by capacitor C'l, as shown in Fig. 30. The 
cffltoring of the charge on C2 maintains the original triggering 
point very closely and results in extended range of the 
control settii^. This triac circuit can be turned on for very 
low levels of applied voltage and is not prone to "extinguish- 
ing" for line-voltf^ drops. 

Fig. 30 Charging cycle of ^jacitoMSac network m Fig. 29 

(reduced hysteresis). 

Because Iriac switching from the high-impedance to the 
low-impedance state can occur in less tlian one microsecond, 
the current applied to the load increases from essentially zero 
to a magnitude limited by the load impedance within the 
triac switching time. This rapid rise of load current produces 
radio- frequency interference (RFI) extending into the range 
of several megahertz. Although this rapid rise does not affect 
television and FM ladici frequencies, it does affect the 
sliorl-wave and AM radio bands. The level of RFI generated 
is well below that caused by small ac/dc brush-type motors, 
but some means of RFI suppression is generally required if 


the triac phase-control circuit is to be used for any extended 
period of time in an ennronmeat in which RFl generation 
cannot be tolerated. 

A rc;is»iiuihly effective suppression technique is shown in Fig. 
31. An inductor Is connected In series with the triac control 
circuit to restrict the current rate of rise, and a filter 
capacitor is used in parallel with the entire network to bypass 
llieb-frequ«nGy s^^ks. 


Fig.31 - RF/-sufif$iiessi0i9 m^verk. 

TIic values shown in Fiji. -^^1 :nc effective in reducing RFl 
noise for mis load currents up to 6 aniperes to such an extent 
that the effects on short-wave and AM signals are either 
nunimized or considered tolerable. For values above 6 
amperes rms, additional suppression can be achieved by use 
of dual chokes in the ac lines to the triac network. 
Depending on the circuit performance required, such 
oppression ntuy or may not be elective and other means of 
tn«; control may be required. 

An alternate method of providing high-current heating 
controls is through use of a proportional c<mtrol circuit using 
int^ral-cycle synchronous switching or zero-voltage switch- 
ir^ This approach varies the aver^ power to the load 
throti^ controlled bursts of full cyd^ of ac voltage to the 


load by turning on the triac at the beginning of the 
mro-voJt^ cr(^i^ Because the triac turns on zeroy 
current, the sudden current steps associated with ph^ 

control circuits and the RFl generated are minimized. The 
RCA-CA3059 zero-voltage switch is a monolithic integrated 
circuit used primarily as a trigger-current generator for con- 
trol of thyristor turn-on during ihe zero-voltage transition. 
This circuit has many features, one of which is a fail-safe 
circuit viiicb inhibits output pulses in event t^t the 
external smsor is opened or diorted. 


This Note has reviewed lliyristors from the viewpoints of 
tesqwrature and voliatic ^.•u[lltlllo^l^. gate trigger eluiracter- 
istics, and effects of SCR's and triacs on circuit performance. 
The availability of power thyristors ^ves deagn e^^ne^ 
greater freedom in achieving circuit simplicity, low cost, and 
small package assembly than electromechanical or tube 
counterparts. Technological improvements are far from 
reaching the saturation level, but are opening new doors for 
circuit applicalion. The impact of Ihyristor Ltppliculions Is 
being felt in normal everyday environments such as 
residential lamp dimming, TV deflection systems, home 
appliances, marine ignition, automotive applications, electric 
heating, comfort controls, and igniters for fuel-fired furnaces. 
Industrial applicaticms for multiple-horsepower motors, lamp 
display boards, inverters, relay protection or replacement, 
radar, sonar, and emergency standby generating systems are 
now finding widespread acceptance in thyrisior controls. The 
introduction of RCA triacs fully characterized tor -4(}0-Hz 
commuiating capability opens the doors to many aircraft 
support api^cations which previoudy were devoid of the 
advantages offered in solid-estate d^gn. It appears that the 
answer to most power-^ntrol applications may be the 

When incorporating RCA Solid State Device* in 
equipment, it it recommended that the designer 
refer to "Operating Considerations tor flCA Solid 
Sate Oevtces", Form No. 1CE-402. available on 
request from RCA Solid State Divi)jort, Box 3200. 
SoRiMYilla. N. J. 08876.